Methods of writing and detecting dibit servo encoding

ABSTRACT

Methods for writing and detecting servo positioning dibit transitions on a magnetic data surface having circular tracks with two magnetic polarizations, the transitions each having a radial extent greater than the spacing between adjacent centerlines. The transitions are written by first prepolarizing the magnetic surface to a first polarization and then writing a second polarization followed by a first polarization and not writing when the write head passes a previously written transition in an adjacent track. The transitions are detected by qualifying the detection means when the sensed servo signal exceeds a threshold amplitude corresponding to a first polarization and detecting the crossing by the sensed servo signal of a second threshold corresponding to the transition to a second polarization.

CROSS REFERENCE TO RELATED PATENT APPLICATION

This invention is a divisional application of prior pending applicationSer. No. 07/490,504, filed on Feb. 28, 1990, now U.S. Pat. No.5,109,307, entitled CONTINUOUS-PLUS-EMBEDDED SERVO DATA POSITION CONTROLSYSTEM FOR MAGNETIC DISK DEVICE, which is a continuation application ofprior application Ser. No. 07/308,963, filed on Feb. 10, 1989, nowabandoned, which was a continuation of prior application Ser. No.07/106,017, filed on Oct. 1, 1987, now abandoned, which was acontinuation of prior application Ser. No. 06/926,885, filed on Nov. 6,1986, now abandoned, which was a continuation of prior application Ser.No. 06/376,971, filed on May 10, 1982, now abandoned.

This invention is also related to commonly assigned U.S. Pat. No.4,536,809, titled ADAPTIVE MISPOSITION CORRECTING METHOD AND APPARATUSFOR MAGNETIC DISK SERVO SYSTEM, issued on Aug. 20, 1985, by the sameinventor hereof, the application having been incorporated by referencein pending application Ser. No. 07/490,504.

BACKGROUND OF THE INVENTION

A magnetic disk device, commonly termed a "disk drive", is a storagedevice used in data processing system for storing retrievable digitaldata in magnetic form. The data is stored on a rotating magnetic disk ina set of concentric circular patterns termed "tracks". A read/write headis mounted on a carriage that moves the head radially to bring it to adesired track and then maintains it in position over that track so thatthe head can record a series of data bits or, alternatively, retrieve aseries of bits from the track as the latter rotates under the head.Large-capacity disk drives incorporate a plurality of such disks mountedfor rotation together on a single spindle. A least one separateread/write head is used for each disk surface, all the heads beingmounted on the same carriage to provide a comb-like arrangement in whichthe heads move in and out in unison.

The carriage on which the read/write heads are mounted is incorporatedin a servo system which performs two distinguishable functions in movingthe carriage. The first of these is a "seek" or "access" function inwhich the servo system moves a read/write head to a selected track froma previous track which may be a substantial number of tracks distant.When the head reaches the desired track, the servo system commences a"track following" function in which it accurately positions the headover the centerline of the selected track and maintains it in thatposition as successive portions of the track pass by the head.

The seek and track-following functions impose different constraints onthe servo system. During a seek operation the carriage must be moved asfast as possible so as to minimize the time required for that operation.Velocity accuracy is also important in establishing a velocitytrajectory and good arrival characteristics. During a track-followingoperation, on the other hand, position accuracy is a most importantfactor. The accuracy with which the read/write head can be made tofollow the track centerline is a determining factor for the trackdensity on the disk. That is, the closer the head can be made to followtrack centerlines, the closer together can the tracks be spaced.

The head-positioning servo system senses the position of the read/writehead by means of servo signals recorded in tracks on the disk pack. Inone conventional arrangement the servo signals are recorded on adedicated servo surface, i.e., a surface which contains only thesesignals. In another conventional arrangement the servo signals areembedded in the data. That is, they are recorded in servo fields at thebeginnings of the data track sectors. The embedded servo signals havethe capability of providing more accurate data head position informationthan the dedicated servo signals. However, because they are spaced apartby the data sectors on the data tracks and are thus sampled periodicallyat a relatively low rate, they are incapable of providing positionsignals having high frequency components. On a dedicated servo surface,on the other hand, the servo signals on each track are essentiallycontinuous and thus they can provide position information having asubstantially broader frequency band.

DESCRIPTION OF PRIOR ART

U.S. Pat. No. 4,115,823, issued to Commander, et al., and U.S. Pat. No.4,072,990, issued to Case, et al., describe a servo control system usingboth embedded servo signals and signals from a dedicated servo surface.The signals from the dedicated surface are used during track seekingoperations. During track following operations, the system combines thesignals from both sources into a "hybrid" position error signal that isused to control the position of a read/write head. Specifically, thehybrid signal is a combination of the low frequency components of theembedded servo signal with the high frequency components of thededicated servo signal. This allows a wide-band hybrid position errorsignal while maintaining the accuracy provided by the low frequencycomponents of the embedded signals. Still, this arrangement does notprovide the positioning accuracy needed for the increased track densitydesired in newer, high performance disk systems.

SUMMARY OF THE INVENTION

The principal object of the invention is to provide a multi-platter diskdrive having an improved head positioning system capable of positioningthe data heads with relatively small error.

A more specific object is to provide a disk drive in which the headpositioning system makes use of both dedicated and embedded servo data.

Another object of the invention is to provide a disk drive of the abovetype capable of closely following data track centerlines and thereforecapable of a relatively high track density.

A further object of the invention is to provide a disk drive of theabove type in which the servo data can be recorded at relatively lowcost.

The disk described herein incorporates a number of features whichaccomplish the foregoing objects. First, instead of combining only thelow frequency components of the embedded servo signal with the highfrequency components of the dedicated servo signal, I combine the entireembedded servo signal with the high frequency components of thededicated servo signal to provide the hybrid or composite track errorsignal used in the servo system. As will be seen from the descriptionbelow, this means that the composite error signal is comprised almostentirely of the embedded servo signal and, thus, is capable of followingthe data tracks more closely than the prior hybrid systems.

As a further feature of the invention, I use high frequency bursts asthe embedded servo signals. Specifically, each sector on a data trackbegins with an embedded servo field containing first a high frequencyburst bordering the track centerline on one side and then a second highfrequency burst bordering the centerline on the opposite side. Thus, asthe servo field passes under the read/write head, the head senses firstone burst and then the other. If the head is displaced from the desiredon-centerline position, the received amplitude of one burst will begrater than that of the other and it is this difference that is used asthe embedded servo position error signal.

This type of embedded servo signal has been used before, but it has not,to my knowledge, been used in conjunction with a dedicated servosurface. It provides an important advantage resulting from the fact thatthe circumferential positions of the servo bursts need not be maintainedwith in tight tolerances as discussed below. Thus, the embedded servosignals can be written on the data surfaces by the same disk drive thatreads and writes data on these surfaces. That is, a disk assembly with apre-recorded dedicated servo surface can be mounted on the drive and thedrive can then use the servo information from the dedicated disk toradially position the heads on the data surfaces and time the Writing ofthe embedded servo bursts. With other types of embedded servo formats,the servo signals must be circumferentially located with a higher degreeof accuracy and fluttering of the data heads relative to the servo headprevents the attainment of such accuracy. Accordingly, both thededicated and embedded servo information must be pre-recorded on ahighly expensive machine designed for that purpose.

Another feature of the drive is the use of novel automatic gain controlcircuits, described below, in the detection of the servo signals. Thesegain control circuits provide wide band widths in the gain control loopsand thus high-gain loops can be used for close following of variationsin the strengths of the recorded servo signals.

Still another feature is the use of a Gray code arrangement in detectingthe passing of track boundaries during seek operations foridentification of the tracks passing the read/write heads. The systemincludes circuitry which makes use of the Gray code in preventing signaljitter from providing false track identifications.

The drive also includes a number of other features which are describedbelow.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1A and 1B contain a diagram of a disk drive embodying theinvention.

FIG. 2A schematically depicts the arrangement of signals on thededicated servo surface.

FIG. 2B schematically depicts the arrangement of embedded servo signalson a data surface.

FIGS. 3A and 3B represent signals derived from the dedicated servosurface as a function of the radial position of the servo head.

FIG. 3C graphically illustrates the derivation of the Gray coderepresentation of track identification.

FIG. 4 is a schematic diagram of the track identification unit of FIG.1B.

FIG. 5 is a table of logic relationships governing the output of thetrack identification unit depicted in FIG. 4.

FIGS. 6A and 6B contain a circuit diagram of the embedded servo datademodulator of FIG. IB.

FIG. 7 is a diagram of various timing relationships in the circuit ofFIGS. 6A and 6B.

FIG. 8 is a circuit diagram of the dedicated servo data demodulator ofFIG. 1A.

FIG. 9 is a circuit diagram of a portion of the automatic gaincontrol/circuit used in the demodulator of FIG. 8.

FIG. 10 is a diagram of the phase-locked loop unit of FIG. 1B.

FIG. 11 is a circuit diagram of the servo signal detector used in thephase-locked loop unit.

FIG. 12 is a circuit diagram of a SYNC signal detector used in thephase-locked loop unit.

FIG. 13 is a diagram of a synchronization pattern detector used in thephase-locked loop unit.

FIG. 14 is a circuit diagram of the velocity estimator of FIG. 1A.

FIG. 15 is a circuit diagram of the position error estimator of FIG. 1A.

FIGS. 16A, 16B and 16C are graphic representations of frequencycharacteristics associated with the position error estimator.

FIG. 17 is a circuit diagram of the dedicated servo offset measurementunit of FIG. 1A.

DESCRIPTION OF AN ILLUSTRATIVE EMBODIMENT

FIGS. 1A and 1B depict in block diagram form a disk drive incorporatinga head positioning system embodying the invention. A mutli-platter diskassembly 10 comprises a plurality of stacked magnetic disks 12, 14 and16 which are mounted in spaced relation for rotation with a spindle 18.A movable carriage 20 supports a set of read/write heads or transducers22-26, positioned to read and write data from the upper and lowersurfaces of the disks 12 and 14 and the upper surface of the disk 16.The carriage also supports a read-only head 27 positioned to read servoinformation recorded on the bottom surface of the disk 16, this surfacebeing a dedicated servo surface.

The carriage 20 is moved to and fro by an electromagnetic actuator 28 todisplace the respective transducers in and out radially with respect tothe disks 12, 14 and 16 so as to access selected circular tracks ofmagnetically recorded information on the disks. The tracks are selectedunder the overall control of a drive control unit 30 that receivesinstructions from a data processing system to which the depicted disksystem is connected. Typically the drive control unit will receiveinstructions to read or write data on a selected track on a selecteddata surface of the disk assembly 20. The present invention is directedto the servo system that moves a read/write head to a selected track andmaintains it in position over the center line of that track during areading or writing operation.

More specifically, with further reference to FIG. 1B, the datatransducers 22-26 are connected to a head selection and amplifying unit32 by a set of conductors 34. The unit 32 selects a data surface in thedisk assembly 10 by connecting to one of the data heads 22-26 inresponse to head select signals from the drive control unit 10. The unit32 is also connected to read/write circuits 34 which convey data fromthe disk assembly 10 to the data processing system during readoperations and in the opposite direction during write operations. Thecircuits 34 also provide an automatic gain control signal for the unit32. During both read and write operations, embedded servo signals areread and write operations, embedded servo signals are read from theselected data track and they are passed by the unit 32 to an embeddedservo data demodulator 36.

At the same time the output of the servo transducer 27 is applied to aconventional preamplifier 38 whose output in turn is fed to a dedicatedservo data demodulator 40. The output of the demodulator 40 appears inan essentially time multiplex manner on four conductors 42A-42D. Thesededicated track position signals are passed to a position estimator 44along with an embedded track error signal from the demodulator 36. Theoutput of the position estimator 44 is a composite track error(COMPOSITE TE) signal which, after modifications described herein, isapplied to a power amplifier 46 which controls the actuator 28, therebyto close the servo loop and thus bring a data head to the desired trackand maintain it on the centerline of that track.

More particularly, the composite track error signal from the positionestimator 44 is summed in a summer 48 with the output of a low frequencygain boost unit 50 and with a misposition correction signal from adigital/analog converter 52. The input to the converter 52 is a digitalrunout and bias force correction signal derived by the drive controlunit 30 in a manner to be described below. The output of the summer 48in turn is passed through a mode switch 54 to a second summer 56. Thesummer 56 adds to the track error signal a velocity estimator 58 and avelocity command derived by a digital/analog converter 60 from signalsprovided by the drive control unit 30.

The output of the summer 56 is passed through a limiting amplifier 62and a frequency compensation network 64 before being applied to thepower amplifier 46.

During seek operations the switch 54 is open so that the input to thesummer 56 consists only of (1) the velocity command signal from thedrive control unit 30, and (2) the velocity feedback signal from thevelocity estimator 58 which is subtracted from the velocity commandsignal by the summer 56 to provide a velocity error feedback signal.

The drive control unit 30 provides the velocity command, which is basedin a well-known manner on the distance of the servo transducer from thetrack to which it is being moved. This distance, which may be termedcoarse position information is provided by a track difference downcounter 66 which is initially loaded by the drive control unit 30 withthe number of tracks to be crossed in moving to the selected track. Thecounter then counts down in response to a track crossing pulse emittedby a track identification unit 68 each time the servo head 27 passesover a track on the way to the destination track. The trackidentification unit 68 each time the servo head 27 passes over a trackon the way to the destination track. The track identification unit 68 inturn responds to output signals from the dedicated servo datademodulator 40 on conductors 42B-42D.

When the selected track is approached, the switch 54 is closed and itremains closed during the subsequent track-following operation, so thatthe summer 56 receives the composite track error signal from theposition estimator 44 as modified in the summer 48. During this mode ofoperation the digital/analog converter 60 receives no velocity commandfrom the drive control unit 30. However, a velocity feedback signal isprovided by the estimator 58.

The velocity estimator 58 derives its velocity signals from thededicated servo position signals on conductors 42A-42D during seekoperations and from the composite track error signal provided by theposition estimator 44 during the track following operations. It isswitched between these two modes of operation by a control signal fromthe drive control unit 30 a indicated in FIG. 1A.

Timing signals for the drive are derived by a phase-locked loop unit 70described below. In particular, the unit 70 provides appropriate timingsignals for operation of the servo demodulators 36 and 40.

FIG. 2A depicts the configuration of the flux transitions recorded onthe dedicated servo surface of the disk 16 (FIG. IB). The servoinformation is recorded in tracks whose centerlines are represented byarcuate lines. The servo information is read by the servo head 27, whichhas an effective width equal to two track widths. The servo surface ispreferably polarized in one direction, for example right to left in thedrawing, and the servo signals are recorded as dibits, each of whichcomprises a first transition from right-to-left polarization to a leftto right polarization and a second transition returning the polarizationto right-to-left. These dibits are depicted with "+/+" legendsindicating the generation of positive pulses in the head 27, followed by"-/-" transitions indicating the generation of negative pulses.

The servo signals in FIG. 2A are arranged in frames, each of whichcomprises a synchronizing field 74 containing synchronizing dibits S1and S2, followed by a positioning field 75 in which the dibits ararranged in radially extending groups designated as A, B, C, and D. Thesignals generated by the head 27 in reading these positioning dibits areused by the system to determine the radial position of the head.

More specifically, in the circumferential positions B and D thepositioning dibits are centered about alternate even-numbered trackswith the B dibits centered, for example, on tracks 78 and 82 and the Ddibits centered on tracks 80 and 84. In similar fashion the A and Cdibits are centered on alternate odd-numbered tracks 77, 79, 81, and 83.Each dibit has a width equal to two track widths. Dibit patterns of thistype are disclosed in IBM Technical Disclosure Bulletin, Vol. 18, No.10, March, 1976, by R. K. Oswald and in U.S. Pat. No. 4,238,809, issuedto Fujiki, et al.

In the system described herein, there is an S2 dibit 30 in everysynchronizing field 74. However, in each track the S1 dibit is omittedin a pattern of frames to provide an index mark. Additionally, there arecontinuous patterns of missing S1 dibits in inner and outer guard bands(not shown). These patterns are not a part of this invention. However,in the embodiment described herein the S1 dibit is never omitted fromtwo successive frames, a factor which will be discussed in connectionwith the description of the phase-locked loop system hereinbelow.

FIG. 3A illustrates the variations in the amplitude of a dedicated servodibit signals as a function of the radial displacement of the servo head27. In this connection it is assumed that in FIG. 2A the amplitude of adibit signal generated by the head 27 is essentially proportional to theportion of the width of head 27 that traverses the dibit. Thus, in theillustrated position of the head 27, the A dibits will generate a signalof full amplitude, whereas B and D dibits will generate half-amplitude,signals and the signal generated by the C dibits will have zeroamplitude. Moreover, when the head is centered on even-numbered trackcenterlines such as centerlines 78 and 80, the A and C dibit signals areboth of half amplitude and, thus, equal. Conversely, when the head 27 iscentered on odd-numbered track centerlines, the B and D dibit signalsare half amplitude and thus equal.

Returning to FIG. 3A, it will thus be apparent that the even-numberedservo track centerline positions of the servo head 27 are a designatedas E, while the odd numbered track positions are designated as 0. Thesepositions are determined by subtracting the A and C signals, and alsothe B and D signals, to ascertain the equalities of the respectiveamplitudes. Subtraction also provides further track information as willnow be described.

With reference to FIG. 3B, the track identification unit 68 makes use ofthe (A-C), (B-D), (D-B), and (C-A) signals from the dedicated servodemodulator 40 (FIG. 1). The (A-C) and (B-D) signals have zero valuescorresponding to centerline positions on even- and odd-numbered tracksas described above. Additionally, the unit 68 is arranged to provide anunambiguous indication of head position within four consecutive trackswhich, for convenience, are numbered as indicated at the bottom of FIG.3B.

Furthermore, as will be understood from FIG. 3C, comparison of thevalues (B+C) and (A+D) and further comparison of the values of (C+D) and(A+B) provide a Gray code representation of these track numbers. Asindicated in FIG. 3C, the transitions in the two Gray code signals arecharacterized by a degree of uncertainty in that when two comparedquantities are approximately equal, noise can result in a jitter of thecomparison signal, i.e., cause it to shift back and forth between onelogic level and the other. The track identification system eliminatesthe effect of this jitter, thus preventing it from adversely affectingoperation of circuits which depend on continuous progression of thetrack identification signals in the proper order.

Turning now to FIG. 4, the track identification unit 68 includes acomparator 90 having an input terminal at which the (A-C) and (D-B)voltages are summed. The comparator 90 thus compares the voltages (A+D)and (B+C) and provides an assertion level output when (B+C) exceeds(A+D). Similarly, a comparator 92 receives the (A-C) and (B-D) voltagesand provides an assertion level output when (C+D) exceeds (A+B). Thusthe outputs of the comparators 90 and 92 correspond to the two Gray codebits of FIG. 3C, whose transitions correspond in turn to the crossingsof the data track boundaries. These outputs are applied to a finitestate machine governed by the table in FIG. 5. This table in turnrepresents successive values in the Gray code position representation ofFIG. 3C.

Specifically, the unit 68 includes a decoder 94, whose input is thecombination of (1) the outputs of a pair of flip-flops 96 and 98representing the Gray code representation of the present track positionof the data heads, (2) the outputs of a pair of flip-flops 100 and 102containing successive samples of the outputs of the comparators 90 and92, and (3) a fifth bit representing the direction of head movement,i.e., forwarded or reverse, as indicated by a signal from a drivecontrol unit 30. The circuit is clocked by a high frequency squarewaveprovided by the control unit 30.

Assume that initially a valid next state signal from the decoder 94 isnot asserted and that a flip-flop 104 is in the reset condition.Successive negative transitions of the clock signals, as passed by aninverter 106, cause the flip-flops 100 and 102 to continually sample theoutputs of the comparators 90 and 92. As long as the flip-flops do notcontain the next valid Gray code representation of the head position,the decoder 94 will continue to negate the valid next state signal andthe unit 68 will essentially idle. When the comparator outputsultimately correspond to the Gray code representation of the next datatrack in the sequence, the five-bit address applied to the decoder 94causes the memory to emit a valid next state signal. This signalconditions the flip-flop 104 to be set by the succeeding positive-goingtransition of the clock signal. In turn the change of state of theflip-flop 104 clocks the states of the flip-flops 100 and 102 into theflip-flops 96 and 98. With the new Gray code state contained inflip-flops 96 and 98, the flip-flops 100 and 102 no longer contain thebits for the next state and the decoder 94 therefore negates the validnext state signal, thereby immediately resetting the flip-flop 104.

The foregoing sequence is repeated each time the outputs of thecomparators 90 and 92 correspond to the Gray code representation of thenext track in the sequence represented by FIG. 3B. On the other hand,assume that after a transition to a new Gray code state, with transferof the new state to the flip-flops 96 and 98, the output of thecomparator 90 or 92 whose transition resulted in the change of statereverses its state in response to noise. This will not cause a change ofstate of the flip-flops 96 and 98, since the false change of stateindicated to the flip-flops 100 and 102 will not be recognized as avalid next state by the decoder 94. Thus, the circuit of FIG. 4 providesan orderly progression of states representing continuous movement of theselected data head from one track to the next during a seek operation.

The decoder 94 transmits to the control unit 30 (FIG. 1B) a binary trackidentification derived from the states of the "present state" flip-flops96 and 98. The decoder 94 also provides a set of track-type signals usedby the velocity estimator 58 (FIG. 1). These signals correspond to thetrack identification numbers as indicated. The memory 94 also provides aone-bit signal indicating whether the present track is an even or oddtrack. The track-type signals and the even/odd signal are derived fromthe states of the flip-flops 100 and 102. The signals are thus subjectto variation because of jitter as discussed above. However, in thecircuits that use them, the instantaneous values are important, not theinformation contained in the flip-flops 96 and 98.

With further reference to FIG. 4, a flip-flop 108, which responds to theresetting of the flip-flop 104, provides a single pulse to the trackdifference counter 66 of FIG. 1, corresponding to each crossing of atrack boundary.

FIG. 8 is a diagram of the dedicated servo demodulator 40 of FIG. IB.The servo signals from the dedicated servo surface arrive at thedemodulator 40 in differential form on a pair of conductors 110 and 112,which apply these signals to a controlled gain amplifier 114 describedbelow in detail. The output of the amplifier 114 is passed through a lowpass filter 116 and a buffer amplifier 118 to a set of peak detectors120, one for each of the A, B, C, and D servo signals (FIG. 2A). Thesesignals arrive at the demodulator in a fixed time sequence and they aregated into the corresponding peak detectors 120 by signals from thephase-locked loop unit 70. The outputs of the peak detectors 120 in turnare applied to a set of differencing amplifiers 122 which provide thedepicted difference signals (A-C), etc.

Since the instantaneous values of the output signals of the demodulator40 are used by other circuits in the system, these signals must beimmunized against variations in the underlying A, B, C, and D servovoltages resulting from such factors as anomalies in the magnetic mediumon the dedicated servo surface. With such effects eliminated, thevoltages will then be truly representative of he radial position of theservo head 27 (FIG. 2A). The demodulator 40 incorporates an automaticgain control circuit to accomplish this function.

At the outset one should bear in mind that no one of the A, B, C, and Dvoltages can be used as a feedback signal for gain control, since thesesignals vary with head position. However, from inspection of FIGS. 2Aand 3A, it will be seen that when the position of the servo head 27corresponds to an odd data track, whose centerline might correspond, forexample, to the centerline of the servo track, the sum of the A and Csignals will be independent of the radial position of the head on thattrack. Similarly, when the head 27 position corresponds to even datatrack, whose centerline might correspond, for example, to the servotrack boundary 84 in FIG. 2A, the sum of the B and D voltages will beindependent of the radial position of the head on that track.

Accordingly, as shown in FIG. 8, the A, B, C, and D voltages from thepeak detectors 120 are applied to a set of summing circuits 124 whichprovide (A+C) and (B+D) voltages to an AGC basis selector 126. In theselector 126 these signals pass through a switch 128 and a switch 130.The switch 130 may be considered to be closed for the purposes of thisdescription, with the switch 128 being switched between the even and oddpositions by the even/odd signal from the track identification unit 68.

Thus the appropriate gain control signal is fed back depending onwhether the position of the servo head corresponds to an odd or evendata track. This signal is then applied to a summer 132 for comparisonwith an AGC reference voltage and the resulting error voltage isintegrated by an integrator 134. The output of the integrator 134 inturn is applied to a feedback function generator 136 that supplies thecontrol current to the controlled amplifier 114.

In order to provide the desired degree of immunity from variations insignal strength, the automatic gain control circuit must have a highloop gain. For stability of operations, this required a large loopbandwidth which is constant over the variations in the strength of theincoming servo signals.

I have found that the requisite bandwidth characteristic can be obtainedapplying an inverse exponential relationship between the gain of thecontrolled amplifier 114 and the AGC feedback voltage that controls thatgain. Specifically, the relationship between the gain K of the amplifierand the feedback voltage, V, should have the form

    K=C.sub.1 e.sup.-C2V

where C1 and C2 are constants. In FIG. 8 this relationship is providedby the function generator 136.

FIG. 9 discloses in detail a circuit incorporating the gain controlledamplifier 114, the integrator 134 and the function generator 136. Thedifferential input signals on conductors 110 and 112 are passed throughblocking capacitors 138 and 140 to voltage dividers comprising seriesresistors 142 and 144 and shunt resistances provided by diodes 146 and148 in the form of appropriately connected transistors. From the voltagedividers the signals pass through a second set of blocking capacitors150 and 152 to a difference amplifier 154 whose output is the output ofthe gain controlled amplifier 114 of FIG. 8. The diodes 146 and 148 areprovided with bias currents by a pair of transistors 156 and 158, eachof whose input is the integrated feedback error signal which is appliedas the base-emitter voltage of the transistor.

With further reference to FIG. 9, integration of the gain control errorsignal from by the summer 132 is provided by an amplifier 160 having afeedback circuit comprising the parallel combination of integrationcapacitors 162 and 164 and resistors 166 and 168. The resistors 166 and168 are current-limiting and do not otherwise affect the operation ofthe circuit. The feedback provided by the capacitors 162 and 164 to thesumming circuit 132 provides the requisite integration at the output ofthe amplifier 160, this output being applied to the emitters of thetransistors 156 and 158.

The circuit of FIG. 9 accomplishes the desired exponential gain-errorvoltage relationship as follows. The collector currents of thetransistors 156 and 158 are exponentially related to the base-emittervoltages of the transistors and are thus exponentially related to thegain-control error voltage provided by the integrator 134. Thesecollector currents pass through the diodes 146 and 148, respectively.The diodes in turn have dynamic resistances which are inverselyproportional to the currents through them. The dynamic resistances ofthe diodes are much less than the resistances of the resistors 142 and144. The voltages applied to the amplifier 154 are therefore essentiallyproportional to the diode resistance and thus inversely proportional tothe collector currents of the transistors 156 and 158. The circuit thusprovides the following two relationships:

    I=C.sub.3 e.sup.C4V

where I is the transistor collector current, and

    K=C.sub.5 /I

where K is the gain of he voltage divider provided by the resistors 142and 144 and diodes 146 and 148. The foregoing formulas can be combinedto provide

    K=C.sub.1 e.sup.-C2V

the desired relationship provided by the feedback function generator 136of FIG. 8.

The phase-locked loop unit 70 is broadly illustrated in FIG. 10.Basically the unit locks a voltage-controlled oscillator 170 to the S2synchronizing signals included in every frame on the dedicated servosurface (FIG. 2A). Input signals from the dedicated servo-datademodulator 40 (FIGS. 1 and 8) are applied to a servo signal detector172, whose output is applied to a SYNC 2 detector unit 174. The detectedSYNC 2 signal in turn is used as one input of a phase detector 176 inthe phase-lock loop of the oscillator 170. The phase error output of thedetector 176 is fed to a charge pump 178 whose output is integrated byan integrator 180, the output of the integrator 180 being the controlvoltage for the oscillator 170. The output of the oscillator 170 isdivided in frequency by a counter 182 which, every time it cycles,applies a "frame" pulse to the phase detector 176. In the illustratedsystem the counter 182 divides the frequency of the oscillator 170 by afactor of 64, and thus the oscillator 170 has a frequency 64 times thatof the SYNC 2 pulses. The oscillator output is also used as a writeclock signal for the read/write circuits 34 (FIG. 1).

Continuing in FIG. 10, the frame pulses from the counter 182 are countedby a sector counter 184. The counter 184 provides a sector pulse eachtime it counts a number of frame pulses equal to the length of a sectoron a data disk. Further, it counts the sector pulses. The counter isreset by an index pulse, developed as described below, each time thedisk assembly rotates past the index position. Thus, sector pulse count,which is passed to the control unit 30 (FIG. 1) is thesector-identifying number. The sector pulses are also used to reset acounter 186 which serves as a timer for the embedded servo data asdescribed below. When the counter 186 is reset, which occurs at thebeginning of each data sector, it begins counting high frequency pulsesfrom the counter 182 and continues until the counter reaches its maximumcount, sixteen in the illustrated example, at which time it disablesitself and thus stops counting until the next sector pulse is received.As will be seen, the frequency of the pulses counted by the counter 186is such that the interval during which the counter counts spans theembedded servo field in each sector of the data track one of the datasurfaces.

The contents of the binary counter 182 are applied to a decoder 188,which provides the A, B, C, and D gate signals used by the dedicatedservo demodulator 40 of FIG. 8 as described above. The decoder 188 alsoprovides SYNC 1 and SYNC 2 gate signals. The outputs of the decoder 188are derived from counts in the counter 182 that occur at the appropriatetimes for these various gating signals.

The phase-locked loop unit 70 of FIG. 10 also includes a SYNC 1 patterndetector unit 190 which derives its input from the servo signal detector172. The pattern detector 190 provides the index pulse discussed above,as well as the signals indicating that the servo head 27 is positionedover an outer guard band or an inner guard band. It also providessignals indicating soft errors as discussed below and an unsafecondition of the phase-locked loop, also discussed below.

The servo signal detector 172 is illustrated in FIG. 11. The inputsignals from the demodulator 40 (FIGS. 1 and 8) pass through an ACcoupler 192 which removes any DC bias from these signals, and thenthrough a low pass filter 194 to the inverting input terminal of anamplifier 196. The output of the amplifier 196 is fed back to thenon-inverting input terminal of the amplifier by way of a pair of diodes198 and 200 and a voltage divider comprising resistors 202 and 204. Theoutput of the comparator switches between a relatively high positivevoltage and a very small positive voltage in response to negativevoltages, respectively, from the low pass filter 194.

The detector 174 operates to detect the negative-going zero crossingbetween the two pulses in each of the servo dibits. As described above,each of these dibits provides a positive pulse followed by a negativepulse. The two transitions of each dibit are spaced close enoughtogether so that the two pulses overlap to provide, in essence, acontinuous transition from a positive peak to a succeeding negativepeak. The point at which this transition crosses the zero axis is welldefined as compared with other parts of the dibit and is quiteinsensitive to variations in overall dibit amplitude. The detector 174operates to detect these transitions.

The positive-going transitions in the output of the amplifier 196correspond to these transitions within the servo dibit signals and it isthese transitions to which the SYNC 2 detector 174 and SYNC 1 patterndetector 190 respond. The diodes 198 and 200 become non-conducting andtherefore offset the slight positive voltage from the output of thecomparator 196 at its positive input when it is at the nonasserted leveland thus assure that, with the feedback, the positive-going comparatortransitions occur at zero input voltage.

FIG. 12 depicts the circuit of the SYNC 2 detector 174 of FIG. 10. Atthe outset, certain characteristics of the dedicated servo signalsshould be kept in mind. First, most frames contain both the S1 and S2dibits. Secondly, the S1 dibit is never absent from two or moreconsecutive frames. Furthermore, in frames containing both S1 and S2dibits, these two dibits are substantially more closely spaced than areany other pair of dibits. Finally, in accordance with the operation ofthe servo signal detector 172 just described, the detection of a dibitis represented by a rising transition of the output of that detector.Similarly, in the output of the SYNC 2 detector 174, rising transitionsrepresent the occurrence of the S2 dibits. The triggered and clockedcircuit elements in the illustrated circuits respond to the risingtransitions of the signals applied to them. Such transitions in theoutput of the detector 172 are referred to as "signals".

Each SYNC signal received from the servo signal detector 172 is appliedto a retriggerable one-shot 210 whose output is passed by a gate 212 andan OR circuit 214 to enable a flip-flop 216. The one-shot 210 assertsits output for an interval slightly longer than the interval between anS1 dibit and the succeeding S2 dibit, e.g., 375 ns. Thus, if the signalthat triggered the one-shot 210 was an S1 signal, the following S2pulse, serving as a clock input for the flip-flop 216 will cause theflip-flop to set, thereby presenting a SYNC 2 signal at its outputterminal. The one-shot 210 will then time out before the next servosignal, which will be derived from a positioning dibit (FIG. 2B),thereby disabling the flip-flop 216. The next signal will thus reset theflip-flop, thereby deasserting its output.

If a frame contains no S1 dibit, the S2 signal will arrive with theflip-flop 216 disabled and will therefore not provide a SYNC 2 outputfrom the flip-flop. To recapitulate, the one-shot 210 and associatedcircuitry provide SYNC 2 output signals during servo frames in whichboth the S1 and S2 dibits are present. When the S1 dibit is absent, thecircuitry in the lower part of the detector 174 enables the flip-flop216 to provide the SYNC 2 output.

More specifically, in frames containing both the S1 and S2 dibits, theoutput of the one-shot 210, in response to the S1 signal from detector172, enables a flip-flop 218 to be set by the succeeding S2 signal. Theoutput of the flip-flop 218 in turn triggers a one-shot 220 whichimmediately resets the flip-flop 218, the output of the latter flip-flopthus being a narrow pulse as depicted. The one-shot 220 asserts itsoutput for an interval somewhat less than the interval to the nextpossible S1 pulse. This signal inhibits the gate 212 and thus provides ameasure of noise immunity by preventing the one-shot 210 from enablingthe flip-flop 216 until the time slot for the S1 signal in the nextframe.

The leading edge of the output of the one-shot 220 triggers avoltage-controlled, variable one-shot 222 which times out after aninterval that nominally terminates, in the next servo frame, 150 nsbefore the arrival of the next S2 signal. At the end of this interval,the rising edge of the output of the one-shot 222 clocks a flip-flop224. If an S1 signal occurs in the new frame before the S2 signal, theresulting output of the one-shot 210 will hold the flip-flop 224 in thereset state. However, if there is no S1 signal in this frame, theone-shot 210 will not have asserted its output at this time. Theclocking of the flip-flop 224 by the output of the one-shot 222 willtherefore set the flip-flop 224, thereby enabling the flip-flop 216 byway of the OR circuit 214. The arrival of the S2 signal in that framewill then set the flip-flop 216, with a resulting SYNC 2 output asdescribed above. The flip-flop 224 will then be reset by the one-shot210 upon the receipt of the next servo pulse from the servo signaldetector 172.

With further reference to FIG. 12 the SYNC 2 detector 174 also includesa phase-locked loop that controls the timing interval of the one-shot222. The rising edge of the output of the one-shot 222 triggers one-shot226 having a timing interval of 150 ns. On termination of this interval,the output of the one-shot clocks an enabled flip-flop 228 and therebysets the flip-flop. The resulting output of the flip-flop 228 enables aflip-flop 230. The flip-flop 230 is clocked by the next pulse from theflip-flop 218.

Assume first that the interval of the one-shot 222 is slightly shorterthan the nominal value, so that it times out somewhat before the 150 nsinterval preceding the S2 signal. The flip-flop 22 will then time outshortly before the S2 signal so that when the latter signal arrives, theresulting clocking of the flip-flop 230 will occur after the flip-flophas been enabled, thereby setting the flip-flop. The resulting output ofthe flip-flop 230 will be passed by a gate 232 to a charge pump 234awhich, for example, applies a positive charge to an integrator 236 whoseoutput voltage controls the timing interval of the one-shot 222. Theintegrator output will change in a direction to increase the timinginterval of the one-shot 222 The interval will then be slightly longerthan the nominal value. The resulting delay in the triggering of theone-shot 226 will delay the setting of the flip-flop 228 and theenabling of the flip-flop 230 until after the arrival of the next S2signal.

The pulse from the flip-flop 218 corresponding to the S2 signal willtherefore arrive with the flip-flop 230 disabled, thereby resetting theflip-flop. The resulting output of the flip-flop 230 will be passed by agate 238 to a charge pump 234b which applies a negative charge to theintegrator 236. The integrator output will thus change in the directionthat shortens the timing interval of the one-shot 222. Thus the timingof the one-shot 222 is "dithered" back and forth in a small intervalencompassing a time corresponding to the interval of the one-shot 256(150 ns) prior to the occurrence of each S2 pulse.

The flip-flop 228 is reset by the output pulses from the flip-flop 218shortly after clocking of the flip-flop 230, as indicated by the delayelement 239. This readies the flip-flop 228 for setting by the output ofone-shot 226 in each cycle of the phase-locked loop.

The above-described operation of the phase-locked loop in the SYNC 2detector 174 requires the receipt of a series of S1, S2 signal pairs.Specifically, if an S1 signal is absent in a frame, there will be nopulse from the flip-flop 218 to clock the flip-flop 230. The latterflip-flop may therefore remain too long in one state, thereby undulychanging the timing interval of the one-shot 222. I therefore employ asignal pair detector that disables the gates 232 and 238 in thosesituations.

Specifically, the output of the one-shot 220, in response to receipt ofan S1, S2 signal pair, triggers a retriggerable one-shot 240 whosetiming interval is slightly longer than one frame. If then a second S1,S2 signal pair is received in the next frame, the output of theflip-flop 218 will clock a flip-flop 242 that is enabled by the one-shot240, thereby setting the flip-flop 242 and enabling the gates 232 and238 to permit operation of the phase-locked loop. If the second SI, S2signal pair is not received, the one-shot 240 to time out, therebydisabling the flip-flop 242. Upon receipt of an S1, S2 pulse pair in thenext frame, a clocking pulse from the flip-flop 218 will arrive at theflip-flop 242 slightly before the triggering of the one-shot 240 by theoutput of the one-shot 220. The flip-flop 242 will therefore remaindisabled.

Finally, if consecutive S1, S2 signal pairs have arrived, with aresulting enabling of the gates 232 and 238 by the output of theflip-flop 242, a subsequent omission of an S1 signal will permit theone-shot 240 will time out, thereby resetting the flip-flop 242 andshutting down the phase-locked loop. Accordingly, the phase-locked loopoperates only after an S1,S2 signal pair that follows an S1,S2 signalpair in the immediately preceding frame.

The above-described operation of the SYNC 2 detector is especiallyimportant in guard band areas, where a significant percentage of SYNC 1signals will be missing.

FIG. 13 depicts the SYNC 1 pattern detector 190. It includes a flip-flop244 which is enabled by the SYNC 1 gate signals generated by the decoder188 in FIG. 10. These gate signals occur during an interval encompassingthe time of arrival of an S1 signal if that signal is present in a servoframe. The flip-flop 244 is clocked by the detected servo signals fromthe detector 172 of FIG. 11. The flip-flop is thus set each time an S1signal is received.

The output of the flip-flop 244 is applied to a shift register 246 whichis clocked by the trailing edge of the SYNC 1 gate signal. Thus, theshift register 246 contains a running pattern of bits representing thepresence or absence of the S1 dibit in successive frames in the servosurface depicted in FIG. 2A. The contents of the shift register 246 areapplied parallel to a latch and decoder 248. The latch and decoder 248latches in the contents of the shift register 246 in response to the Agate signal, thus being the first signal in each frame generated by thedecoder 188 of FIG. 10 following the receipt of the S signal.

The latch and decoder 248 provides an output signal if the bit patterncontained therein indicates any of the conditions indicated in FIG. 13.The generation of the index, outer guard band and inner guard bandsignals have been discussed above. The soft error signal is asserted ifthe bit pattern has a one-bit error. The soft error signal is assertedif the bit pattern has a one-bit error. The PLO UNSAFE signal isasserted if the bit pattern has an error in two or more bits. Thissignal is an indication of the possible unreliability of the outputs ofthe phase-locked loop unit 70.

In FIG. 2B I have illustrated the format of the embedded servoinformation in data tracks on one of the data surfaces in the diskassembly 10. A series of data tracks are denoted by their centerlines250, 251, etc. Each track contains a series of sectors, each of whichincludes a servo field 254 followed by a data field 256. The servofields 254 contain two sets of servo signal blocks, designated "X" and"Y". Each block is centered on a track boundary (not shown), has a widthof one track, and thus spans the distance between the two adjacent trackcenterlines. The X blocks are centered on alternate track boundaries andthe Y blocks are centered on alternate boundaries staggered with respectto the blocks X. Thus, as the disk rotates under a read/write head 262positioned over the track centerline 251, a portion of an X servo blockwill pass under the head followed by a portion of a Y servo block. Thesystem determines the radial position of the head 262 with respect tothe centerline 251 by ascertaining the relative proportions of thewidths of the X and Y blocks passing under the head.

More specifically, each X or Y servo block contains a magneticallyrecorded high frequency burst. The amplitude of the burst received fromthe X block by the head 262 is compared with the amplitude received bythe succeeding Y block. Equality of the two indicates that the head iscentered on the centerline 251; if they are unequal, the magnitude ofthe difference of the detected amplitudes is a measure of the off-centerdistance of the head 262.

It should be noted that when the head 262 is positioned over an oddnumbered track, such as the track 251, the X servo blocks are radiallyouter blocks with respect to the track centerline and the Y blocks areradially inner blocks. Conversely, when the head 262 is positioned overan even numbered track, the Y blocks are outer blocks and the X blocksare inner blocks. This causes the sense of the position error signalderived from the X and Y blocks to depend on whether the track iseven-numbered or odd-numbered, a factor that is taken into account inthe operation of the embedded servo detector 36.

It will be apparent that the X and Y servo blocks in FIG. 2B need not bepositioned in the circumferential direction with a high degree ofaccuracy. They should be spaced apart circumferentially by a distancegreater than the width of the magnetic ga in the head 262 so that thehead will not receive energy from an X block simultaneously with thereceipt of energy from a Y block. Also, there should be deadbands beforethe X blocks and after the Y blocks, so that the head 262 will notreceive any other signals when it is receiving X block or Y blocksignals. Other than that, the circumferential positioning of these servoblocks is subject to a relatively wide tolerance, depending on theconstraints imposed by the timing signals used in detecting the servoinformation.

A resulting important feature of this arrangement is the ability to usethe disk drive itself in recording the embedded servo signals. A highlyaccurate servo recording system need be used only to record the signalson the dedicated servo surface. The signals retrieved from that surfacemay then be used by the illustrated drive itself in recording theembedded servo information on the other data surfaces.

FIG. 6 is a diagram of the detector 36, which detects the embedded servosignals and develops the embedded position error signal in responsethereto. The various timing signals used in FIG. 6 are generated by adecoder 262 whose input is the content of the counter 186 of FIG. 10.The timing of these signals and the signals processed by the demodulator36 are depicted in FIG. 7.

With reference to FIG. 6, the input to the demodulator 36, from the headselection and amplifier unit 32, passe through a summer 264 to an inputamplifier 266. The output of the amplifier 266 is passed through an L-Ctank circuit 268 whose resonant frequency equals the frequency withinthe signal burst contained in each of the X and Y servo data blocks(FIG. 2B). The tank circuit 268 is keyed on and off by a reset L-C tanksignal from the decoder 262, this signal effectively short-circuitingthe tank circuit at times other than the times of arrival of the X and Ybursts, as indicated in FIG. 7. The form of the output of the tankcircuit 268 is also depicted in FIG. 7.

The output of the tank circuit 268 passes through an amplifier 270 to afull wave rectifier 272 whose output is filtered by a low pass filter274. The filtered signal then passes through a switch 276 and aresetable integrator 278, a typical integrator output being depicted inFIG. 7. From the integrator 278, the signals pass through a buffer 280to a pair of sample-and-hold circuits 282 and 284. If the servo burstcontained in the integrator 278 is from an outer block, it is sampledand held in the circuit 282. If it is from an inner block, it is held inthe circuit 284. Thus, at the end of each embedded servo field, thesample and hold circuits 282 and 284 contain updated voltagescorresponding to the sensed amplitudes in the outer and inner servoblocks in the data track on which the selected data head is positioned.

The contents of the sample-and-hold circuits are passed through bufferamplifiers 286 and 288 to an automatic gain control circuit generallyindicated at 290. After gain correction by the circuit 290, they aresubtracted in a differencing amplifier 292 whose output is the embeddedtrack error signal.

More specifically, with further reference to FIG. 6B, in the automaticgain control circuit 290, the outer and inner signals pass throughswitch attenuators 294 and 296, low pass filters 298 and 300, and bufferamplifiers 302 and 304. The outputs of the buffer amplifiers are appliedto the differencing amplifier 292. They are also applied to a summer 306which compares the sum of their voltages with a reference voltage. Avoltage comparator 308 asserts its output whenever the sum of the outputvoltages of the buffer amplifiers 302 and 304 exceeds the referencevoltage. In response to the assertion level, a switch driver 310switches the attenuators 29 and 296 to their attenuating condition. Withthe illustrated attenuators, each of which comprises a series resistorand a shunt switch, this drops the output voltages of the attenuators,to zero. The output voltages of the low pass filters 298 and 300thereupon begin to decrease and when their sum becomes less than thereference voltage applied to the summer 306, the comparator 308deasserts its output. As a result, attenuators 294 and 296 are switchedto their non-attenuating condition so that the output voltages of thelow pass filters 298 and 300 once again begin to increase.

In operation, the attenuators 294 and 296 are rapidly cycled back andforth between their attenuating and nonattenuating conditions, the rateat which this switching occurs depending on the time constant of the lowpass filters 298 and 300 and the dead band of the comparator 308.Preferably the comparator 308 has a negligible dead band and as a resultthe attenuators 294 and 296 are switched at a rapid rate, e.g., 1-5 MHz.The gain control circuit 290 thus has a fast response, typically ofseveral microseconds, to changing input signal conditions andfurthermore can accommodate a wide range of input signal levels.

The circuit of the velocity estimator 58 (FIG. IA) is depicted in FIG.14. During seek operations, the velocity estimator 58 derives itssignals by differentiating the respective position error signalsprovided by the dedicated servo data demodulator 40 (FIG. 1B). As seenin FIG. 3B, the mid-voltage portion of the (A-C) signal occurs when theservo head is over tracks identified as "00". This mid-portion is themost linear part of the signal. Therefore, it is used for velocityestimation and also, as described below, for position estimation. Forthe same reasons, the (D-B) signal is used when the servo head is overan "01" track, the (C-A) signal is used when the head is over a "10"track, and the (B-D) signal is used when the head is over a "11" track.The velocity estimator 58 selects these signals by means of the trackdesignations provided by the track identifying unit 68 (FIG. 1A).

More specifically, returning to FIG. 14, the position signals from thedemodulator 40 are applied to a set of differentiator input sections312, 314, 316, and 318, each of which comprises a series capacitor 320and a series resistor 322, followed by a switch 324. The switches 324are used to selectively connect the input sections 312-318 to adifferentiator output section 326 that comprises an operationalamplifier 326a provided with feedback by means of a resistor 326b. Forvelocity estimation during track following operations, the negativeposition error signal from the position estimator 44 (FIG. IA) isapplied to a differentiator input section 328 which includes, as anadditional element, an inverter 330 because of the inverted nature ofthe input signal to that section.

For the two modes of normal system operation, the velocity estimator isswitched by a dedicated/embedded selection signal from the drive controlunit 30. When this signal is asserted, it enables a set of gates 332-338to pass track-type signals from the identification unit 68. Thesesignals control the switches 324 in the differentiator input sections312-318. Thus, in accordance with the track-type identification, theappropriate input signal from the demodulator 40 is differentiated toprovide the velocity output signal of the estimator 58 When thededicated/embedded signal from the control unit 30 is deasserted, i.e.,during track following operations, the output of an inverter 340 enablesthe switch 324 in the input section 328, so that the velocity output isderived from the composite track error signal.

A feature of the circuit is the configuration of each of the switches324. Each switch is a two-position switch which, when enabled, connectsits differentiator input section to the output section 326. When theswitch is disabled, it is not merely opened, but rather it connects theinput section to ground. Thus the right-hand electrode of the capacitor320 in that section is held at ground potential. Furthermore, the inputterminal of the amplifier 326b is maintained at ground potential byvirtue of the negative feedback provided by the resistor 326b.Accordingly, when one of the switches 324 is enabled, the right-handelectrode of the capacitor 320 in that input section is already at thesame potential as the input terminal of the differentiator outputsection 326. Consequently there is no current surge as a resulting ofthe enabling of any of the switches, with a resultant prevention ofvoltage spikes in the velocity signal.

The velocity estimator 58 also includes a novel arrangement for reducingthe effects of high-frequency noise in the velocity signal. Thedifferentiation that provides the velocity signal also emphasizeshigh-frequency noise components which can have an adverse effect onsystem operation. One might reduce the effect of such noise by passingthe velocity signal through a low-pass filter having an appropriatelyhigh half-power frequency. However, a filter will affect the phase ofsignal components at lower frequencies and in particular at frequencieswithin the pass band of the servo loop, thereby presenting aninstability problem if high loop gain is used. Instead of using aconventional low-pass filter, I reduce the noise by appropriateselection of the conventional slew rate capacitor connected to theamplifier 326a. This capacitor, which is indicated at 326c, has asubstantially higher capacitance than conventional slew rate capacitors,e.g. 30 pf. Slew rate control in a conventional operational amplifierprovides an essentially unaffected gain and phase characteristic out toa slew rate (frequency-amplitude combination) determined by the slewrate capacitor. At that frequency there is a sharp change incharacteristics, with the gain dropping off at a rapid rate. I selectthe capacitance of the capacitor 326c to set the slew rate to correspondto a carriage acceleration slightly greater than the maximumacceleration of which the positioning system is capable. Thissubstantially reduces the noise in the velocity signal while leavingunaffected the gain and phase characteristics of the servo loop.

FIG. 15 depicts the circuit used in the position estimator 44. Theposition error signals from the dedicated servo demodulator 40 arereceived by a set of switches 342 which selectively apply these signalsto a buffer amplifier 344. Each switch 342 is enabled in accordance withthe track-type identification provided by the track identification unit68. This applies the appropriate input signal to the amplifier 344according to the track type over which the selected data head ispositioned (FIG. 3B). The output of the amplifier 344 is applied to aselector switch 346. The same signal is also passed through a high passfilter 348 to a summing junction 350. The embedded track error from thedemodulator 36 is also applied to the summing junction 350, whose outputin turn is passed to the selector switch 346. The signal selected by theswitch 346 is applied to the input terminal of an amplifier 352 providedwith negative feedback as shown. The output of the amplifier 352 is thecomposite track error signal.

The switch 346 operates in response to the dedicated/embedded signalfrom the control unit 30. When this signal is asserted, the switch 346applies only the dedicated track error signal to the amplifier 352. Theresulting composite track error signal is used by the servo system whenthe selected data head is within 2.5 tracks of the selected trackcenterline but is still more than one-half track distant from thecenterline. When the selected head comes within a distance of one-halftrack from its final on-centerline position, the control unit 30deasserts the dedicated/embedded signal so that the switch 346 connectsthe summing junction 350 to the amplifier 352. The composite track erroris then the sum of the embedded track error and the output of the highpass filter 348. In particular, it is the sum of the embedded trackerror signal and the high frequency components of the dedicated trackerror signal. The summed signals have the characteristics indicatedalgebraically in the boxes 348 and 354 in FIG. 15.

The characteristics of these signals are graphically depicted in FIGS.16A and 16B. As shown in FIG. 16A, the frequency characteristics of theembedded position error signal, while exhibiting a high-frequencycut-off at the sampling frequency, f_(s), are not the same as thecharacteristics of a low pass filter.

Specifically, this signal exhibits an appreciable phase lag at afrequency much lower than the amplitude cut-off frequency. As a result,if one is to pass the embedded error signal through a low-pass filterand the dedicated sign;al throug a high-pass filter and then combinethem as is conventionally done, the common half-power frequency of bothfilters has to be unduly low in order to avoid problems resulting fromthe phase characteristic of the embedded error signal. Indeed, in priorsystems, the common half-power frequency is 100 Hz. Since the diskposition servo loop commonly has a band width of the order of 500 Hz,reduction of the embedded error signal at such a low frequency resultsin a loss of much of the highly accurate position information containedin this signal.

I have found that a substantially improved composite error signal can beobtained by not filtering the embedded error signal, as shown in thecircuit of FIG. 15. The dedicated error signal is passed through ahigh-pass filter having a half-power frequency much higher than in priorsystems. For example, I have determined empirically that a ratio ofapproximately 4 to 1 between the sampling frequency of the embeddederror signal and the half-power frequency of the high pass filter 348(FIG. 15) provides a highly accurate, fast responding composite positionerror signal.

For example, assuming that each data track contains 64 sectors and therotational speed of the disks is 3600 rpm, the sampling frequency forthe embedded error signal will be 3840 OHz as indicated in FIG. 16A.With the filter 348 of FIG. 15 having a half-power frequency of 1 KHz,the filtered dedicated error signal will have the frequencycharacteristics exhibited in FIG. 16B. The composite signal will thenhave the characteristics shown in FIG. 16C. As shown therein, thissignal which includes a full contribution of the embedded error positionsignal, has excellent phase and gain characteristics beyond the nominal500 Hz band width limit of the servo loop. Thus the composite positionerror estimator described herein makes full use of the embedded positionerror signal without suffering from loop instability.

The position estimator 44 includes an arrangement, similar to that ofthe velocity estimator 58 (FIG. 14), for preventing voltage spikes whenthe high-pass filter 348 is switched into operation. The filter 348includes a capacitor 357 in series with a resistor 359, a switch 361connects the capacitor to resistor 359 when the embedded/dedicatedsignal is asserted. At other times it connects the capacitor to ground,the potential at the input terminal of the amplifier 352. Thus, when thefilter is connected to the amplifier, it does not cause a current pulsein the amplifier input.

When the system described herein is used to write the embedded servodata onto the data disks, the position error signals must be derivedsolely from the dedicated servo disk. Moreover, as explained above inconnection with the description of FIG. 2B, the embedded servo blocks258 and 260 are displaced one-half track from the centerlines of thedata tracks. That is, they are centered on the boundaries of the datatracks. Turning to FIG. 3B, it will be seen that the (A-C), etc., errorsignals provided by the dedicated servo demodulator 40 of FIG. 1B can beused for such half track positioning. For example, at the boundary Jbetween the "10" and "11" tracks, the sum of the (C-A) and (B-D) signalsis 0. Moreover, the sum increases with head movement in the forwarddirection from that boundary and decreases with head movement in thereverse direction. Accordingly, the sum of these two signals can be usedto position the data heads on the boundaries between "10" and "11" datatracks. Similarly, other additive combinations of the signals depictedin FIG. 3B can be used to position the data heads on other trackboundaries for the writing of embedded servo blocks.

Referring next to FIG. 15, during track-boundary or "half-track"positioning of the data heads, control unit 30 provides selectionsignals to the switches 342 that select pairs of input signals inaccordance with the signal pattern depicted in FIG. 3B. The sum of twoselected signals is therefore applied to the buffer amplifier 344 and,with the dedicated/embedded signal asserted, this sum is used as theoutput from the position estimator 44 to position the data heads. Toaccommodate this type of operation I have included a set ofequal-resistance resistors 356 in series with the switch 342. The bufferamplifier 344 has an essentially infinite input resistance. Accordingly,when a single switch 342 is enabled, the corresponding input signal isreceived by the buffer amplifier 344 without attenuation. On the otherhand, when a pair of input signals is selected, for track-boundaryoperation, the two resistors 356 involved serve as voltage dividerswhich reduce each of the signal amplitudes by half. The sum of the twoamplitudes is thus equal to the amplitude of a single input signal andthe servo loop can operate with the same gain as with normal operation.

As noted above, toward the end of a seek operation, when the servo head27 (FIG. 1) is within 2.5 track widths of the destination track, theservo positioning loop takes over. Specifically, the drive control unit30 closes the position mode switch 54 (FIG. 1B) to apply the compositetrack error signal to the summer 56. Also the dedicated/embedded signalis asserted so that the track error signal consists solely of the outputof the dedicated servo data demodulator 40 (FIG. 1B). Referring to FIG.3B, assume, for example, that the destination track is a "00" trackwhose centerline is at the point indicated at 367. When the servo headis over that track, the (A-C) signal can be used as an position errorsignal. However, assuming that the head is approaching the destinationtrack from the left, this signal cannot be used before the head arrivesat the centerline of the preceding "11" track.

Accordingly, when the head arrives at the boundary K between the tracksdesignated "01" and "10" in FIG. 3B, the drive control unit 30 (FIG. IB)causes the position estimator 44 (FIG. IA) to select the (B-D) signal.At the same time, the drive control unit applies an offset to thedigital/analog converter 60 corresponding to a downward shift of the(B-D) curve by an amount that makes it serve as a linear extension ofthe (A-C) signal at the boundary "L" between the "11" and "00" tracks.Specifically, as can be seen in FIG. 3B, the offset corresponds to adistance of one track. Than when the head reaches the track boundary"L", the drive control unit 30 removes the offset and switches the (A-C)signal to the input terminal of the position estimator 44. The servosystem than continues bringing the head within the destination "00"track. At that point the control unit deasserts the dedicated/embeddedsignal to change the output of the position estimator 44 to acombination of the dedicated and embedded error signals as describedabove.

During track-boundary operations the signals from the demodulator 40used by the position estimator 44 are derived from all four of the servosignals (A, B, C, and D) in the dedicated servo tracks. Accordingly,with reference to FIG. 8, a half-track signal from the control unit 30causes the selector switch 128 to select, for automatic gain controlfeedback, the sum of all of these signals.

The illustrated disk drive also includes provisions for applyingcorrections to the composite position error signals. These correctionsare of two types: one is compensation for various offset or zero shiftfactors in the system. The net effect of these factors is to shift thezero point of the composite track error signal, with the result that thesystem maintains the heads at positions displaced from the desired trackcenterline positions. The second type of correction is of a more dynamicnature. It compensates for such factors as track runout and bias forcesexerted on the carriage 20 (FIG. 1B). These factors are functions of theangular position of the disks and the radial position of the heads,respectively.

Some of the offset errors are corrected by an offset correction voltageinjected into the position estimator 44 (FIG. 15) by an offsetcorrection digital/analog converter 358 (FIG. IA). Others are correctedby adjustment of the digital signals applied to the velocity commanddigital/analog converter 60. The drive control unit 30 of FIG. 1B isprogrammed to run through the following routine to ascertain and applythe various offset correction factors.

First, with reference to FIG. IA, the control unit 30 applies a groundcalibrate signal to a low pass filter 360 which normally receives itsinput signal from the position estimator 44 and applies its outputsignal to an analog/digital converter 362. The ground calibrate signalgrounds the input of the filter 360 and the drive control unit 30 takesin the output of the converted 362. Any output other than zero indicatesan offset within the combination of the filter 360 and converter 362.

The control unit 30 then deasserts the ground calibrate signal andapplies a calibration generation enable signal to the embedded servodata demodulator 36 (FIG. IB). As shown in FIG. 6A, this signal enablesa square wave generator 364 that applies its square wave signal to thesumming junction 264 at the input of the demodulator 36. The headselection unit 32 is shut off and the squarewaves from the generator 364thus provide the only input to the demodulator 36. If there is any zerooffset in the demodulator, it will then appear as an output signal fromthe demodulator 36.

Returning to FIG. 1A, the drive control unit also loads a content ofzero into the digital/analog converter 358 so that any offset in theconverter will be applied to the position estimator 44 along with anyoffset in the embedded servo data demodulator 36. Furthermore, thecontrol unit 30 asserts the dedicated/embedded signal. The compositetrack error from the position estimator 44 thus includes the offsets ofthe demodulator 36 and the digital/analog converter 358. It does notinclude any offsets from the dedicated servo data demodulator 40, sincethose are excluded by the high pass filter 348 (FIG. 15).

The two offsets to be measured are thus passed by the low-pass filter360 to the analog/digital converter 362 and the drive control unit 30then takes in the resulting digital representation of the sum of thosetwo offsets. It modifies the sum in accordance with the previouslymeasured offset in the filter 360 and converter 362. It then uses thisnumber as the basis for adjusting converter 358 and iterates until acorrection value is established that makes the composite TE voltageessentially zero. Finally, it stores the result in a memory 363. Duringtrack-following operations, the control unit 30 applies the result tothe digital/analog converter 358. The converter 358 thus feeds theposition estimator 44 with a correction voltage that compensates for theoffsets in the demodulator 36 and converter 358.

During normal track-following operations of the system, offsets in theoutput of the dedicated servo demodulator 40 are insignificant to systemoperation since they are removed by the high pass filter 348 (FIG. 15).However, the d.c. components in the outputs of the demodulator 40 areused in maintaining the data heads at their track boundary positionswhen the system is used to write the embedded servo signals on the datasurfaces. With reference to FIG. 3B, consider, for example, thegraphical representation of the (A-C) signal as a function of headposition. This signal has a nominal zero value at the track centerlinesdesignated at 365, 366, and 367. In particular, the signal undergoes acomplete cycle as the servo head moves from the centerline 365 to thecenterline 367. If there is a offset in this signal there will be adisplacement of the zero points so that the distance from the zero point365 to the zero point 366 will differ from the distance from the zeropoint 366 to the zero point 367. The system calculates the offset ineffect by measuring these distances.

More specifically to measure the offset in the (A-C) signal, the drivecontrol unit 30 deasserts the dedicated/embedded signal, selects one ofthe input signals, e.g., (A-C), for the position estimator 44 andinitiates a slow speed constant velocity seek operation. At the sametime, it activates a dedicated servo demodulator offset measurement unit368. The circuit for the measurement unit 368 is illustrated in FIG. 17.The control unit 30 asserts a count enable signal that enables aflip-flop 400. The composite track error signal from the positionestimator 44 is applied to a comparator 402 and when the track errorsignal undergoes a positive-going transition, it clocks the flip-flop400 and thereby sets it. This enables a pair of gates 404 and 406. Thegate 406 thus passes the output of the comparator 402 to the enableinput of a counter 408. The counter thereupon begins counting pulsesfrom a high frequency clock. When the position error signal undergoes atransition to the negative portion of its characteristic, the comparator402 deasserts its output, thereby disabling the counter 408. However, byway of an inverter 410, comparator output enables a second counter 412to count the clock pulses. The counter 412 is then disabled during thenext transition of the comparator 402 output. It will then be seen thatthe counter 408 counts during positive portions of the (A-C) signal andthe counter 412 counts during the negative portions. This operationcontinues, with the counters counting the intervals of successivepositive and negative portions of the error signal, a substantial numberof such intervals being counted to reduce the effects of noise. When thecontrol unit 30 determines that the sum of the contents of counters 408and 412 reach a predetermined level it deasserts the count enable signaland the next rising transition from the comparator 402 clocks theflip-flop 400, thereby resetting the flip-flop and disabling the gates404 and 406. With this arrangement, the counters 408 and 412 measure thelengths of an equal number of "half-cycles" of the signal.

The difference in the contents of the counters 408 and 412 is a measureof the offset of the (A-C) error signal. The drive control unit 30calculates an appropriate change in correction value. It then uses thisnumber as the basis for adjusting converter 358 and iterates thisprocedure until a correction value is established that makes theeffective measured offset in the composite TE signal essentially zero.Finally, it stores the correction value in the memory 363. The operationis then repeated for the other output signals from the demodulator 40(FIG. IB). The correction values thus obtained are then sent to theoffset correction digital/analog converter 358 (FIG. IA) during writingof the embedded servo signals. Specifically, the drive control unit 30applies the average of the offset corrections for each pair of signalsfrom the demodulator 40 used by the position error estimator 44 duringservo writing.

Finally, the drive control unit measures the carriage bias force and thetrack runout, the latter measurement being made separately for each diskin the disk assembly 10. The corrections required to compensate forthese offsets are stored in memories 370 and 372 (FIG. IB). The controlunit 30 applies the sum of these corrections to the digital/analogconverter 52 (FIG. 1A) in accordance with the radial position of theheads (bias force) and the angular position of the disks (runout). Whilethere are a number of available methods for ascertaining thesecorrections, I prefer to use the method disclosed in my copendingapplication referenced above.

The invention also relates to the method in which the servo signals arewritten on the dedicated servo surface. The servo head 27 has a width oftwo servo tracks, as explained above. It can thus write an entire A, B,C or D dibit at one time. Assume that the servo surface were to bewritten in a conventional manner beginning at the top of FIG. 2A. Thedata head would switch to a full positive current, then to a fullnegative current in writing the first S1 dibit, the negative currentwould remain on until the beginning of the S2 dibit when it would switchto full positive and then a full negative value to write that dibit. Onarriving at the position of the A dibit, the polarity of the headcurrent would again reverse to positively polarize the magnetic mediumand then switch to a negative value to negatively polarize it. Ignoringfor the purposes of this explanation the presence of a portion of a Ddibit, the head would continue to magnetize the medium in the negativedirection until the position of the next S1 dibit. Thus, in the regionbetween dibits, the medium would be polarized in the negative direction.This background polarization is required to ensure the absence ofspurious signals in reading from the disk.

After completion of the first revolution of the disk surface, the headwould be moved downward (FIG. 2A) by one track to write the B dibits.One way to accomplish this would be to proceed as in the previousrevolution and rewrite the lower halves of the A dibits, with thenegative current in the head remaining until the positions of the Bdibits are reached, then switching the head current to the positive andthen to negative direction to write the B dibits.

This procedure presents two problems. In the first place it isdifficult, if not impossible, to write a new lower half of a dibit thatwill be essentially exactly aligned with the upper half. Thismisalignment is acceptable in the synchronizing dibits, but in thepositioning dibits it adversely affects the position-sensing capabilityof the system. To overcome this problem, one might turn off the writingcurrent before reaching each A dibit and then turn it on again beforereaching the position of the succeeding B dibit. However, this would notsolve the second problem, which would occur during the writing of the Cdibits.

Specifically, when the head is moved to the next track to write the Cdibits, it would have to polarize the medium in the intervals from theS2 dibits to the B dibits. In passing the A dibits, the fringe fieldsfrom the head would change the magnetizations in the lower portions ofthe A dibits. This in turn would alter the characteristics of the signalreceived from the A dibits and, in particular, would adversely affectthe linearity of the received signal as a function of the radialposition.

I have overcome these problems by first pre-polarizing the entire disksurface with the desired background or interbit polarization. In thatcase, when writing the C dibits, for example, the servo head need notpolarize the disk in the region below the A dibits, that polarizationhaving been accomplished before the A dibits were written. Thus the Adibits are unaffected by the writing of the C dibits. To write the Cdibits, for example, the servo head is turned off after the writing ofthe S2 dibit. It is turned on again in the negative (background)direction after passing the B dibits. This causes no change in themagnetic surface since it is already polarized in the same direction.Then, when the position of a C dibit is reached, the polarity isswitched to the positive direction and then back again to the negativedirection to write the dibit. It is left on just long enough to writethe C dibit and is turned off before the D dibit period. With thisprocedure I have obtained the linearity of the position signals derivedfrom the positioning dibits required for close tolerance operation ofthe head-positioning system.

What is claimed as new and desired to be secured by Letters Patent ofthe United States is:
 1. A method for writing servo positioning dibittransitions on a magnetic data surface having circular data tracks withfirst and second magnetic polarizations, said transactions each having aradial extend greater than the spacing between adjacent trackcenterlines, comprising the steps of:A) prepolarizing said magneticsurface to said first polarization; B) positioning the centerline of amagnetic head of width greater than the spacing between adjacent trackcenterlines over the centerline of one of said circular data tracks; andC) writing upon said track positioning dibit transitions by switchingwrite currents on in said magnetic head to provide said secondpolarization followed by said first polarization and switching saidwrite currents off at least when said head passes a previously writtendibit in an adjacent track.